Nixie Tube Clock

Design files for this project can be found here

For the longest time now, I’ve wanted to build myself a Nixie tube clock completely from scratch. Building one isn’t easy though given the various barriers to entry: the high cost of Nixie tubes ($10-$50 each), high voltages for driving the tubes (~170V), designing the PCB, the desire for an nice enclosure that isn’t made out of laser cut acrylic.. All together, it meant that I didn’t have the time and resources to put towards this project until now.

Since I anticipate that I’ll be actively working on this project over the next 8-16 months, this post will not have the usual changelog at the top. Instead, I’ll be slowly appending my progress to this post along with any useful info I find.

Feature Planning

Required Features

  • Nixie tubes, IN-8 for first revision
  • Compact metal enclosure (CNC’d steel/brass)
  • RTC with backup battery
  • Sunrise projection lighting
  • AC powered (wall transformer)
  • Capacitive touch snooze button
  • Tactile configuration buttons (CNC’d)
  • HH:mm (4 tubes + separator)
  • Dimmable brightness

Features Under Consideration

  • GPS for automatically setting time/location
  • Audio via surface transducers on bottom
  • OLED/VFD front display
  • Custom sockets for easy tube replacement
  • Motion/light sensors
  • Current-controlled drivers
  • AM/PM indicator
  • Wireless connectivity (unlikely)
  • Battery powered via 18650 (unlikely)


Nixie tubes come in two orientations, side view (IN-8/IN-14/IN-18) and end view (IN-12/IN-17). A few good sites for comparing different tube models: Dieter’s Nixie World site has both side and end views from a variety of different manufacturers, Steampunk Alchemy’s site has pictures on soviet models and others, and Sphere Research’s Nixie Tube site has pictures on a large number of models as well.

Soviet tubes looks to be widely available on eBay, and is thus likely a better choice since it’ll be cheaper and easier to replace parts. Non-soviet tubes are roughly the same price for similar display sizes but have different digit shapes at the cost of being rarer. A quick comparison of common soviet tubes available on eBay (shipping adds +$10):

IN-17 — side mount, 9mm height, ~$4 each
IN-12 — side mount, 18mm height, ~$1.5 each (IN-12B has a left dot, IN12-A does not)
IN-14 — vertical mount, 18mm height, ~$4 each, long leads, has a fine grid version, pointed taper at the top
IN-8 — vertical mount, 18mm height, ~$10 each, flatter top compared to IN-14
IN-8-2 — same as IN-8 but with longer leads, has an alternate version with a much smaller grid
IN-18 — vertical mount, 40mm height, ~$50 each, huge but expensive

For a clock, you’ll also need a divider of some sort between the digits. Similar to Nixie tubes, there are neon bulbs that operate at a similar voltage with a similar color output. These bulbs are available in both side view (IN-3/IN-29) and end view (IN-6/IN-24/IN-28) variations. Neon bulbs are inexpensive compared to Nixie tubes due to their size, and can be found on Ebay in much larger quantities for cheap (10¢ for an IN-3). Note that TH4B/TH5B/MTH-90 models, while commonly advertised as Nixie tubes, are actually Thyratron tubes used for switching and rectifiers.

Preliminary Testing

For initial testing, a small high voltage power supply with a variable voltage output of 85-235V was used. The supply voltage was set at 170V and a 15k ohm resistor was used to limit the current to a safe level. The voltage drop measured across the resistor came out to be around ~37V, which corresponds to a ~133V drop across the tube and a drive current of ~2.5mA as the Nixie effectively short-circuits once the tube is ignited. The complete turn-off voltage was sightly lower than the voltage drop at ~124V. These values varied slightly between tubes, but not enough to really cause a difference in overall brightness. The ideal values for driving these tubes is unknown since the seller didn’t supply a datasheet, but most resources listed an anode voltage requirement of 170-200V and an ideal cathode current of 2.0-3.5mA for IN-8 tubes.

The interesting observation here is that the tubes exhibits a large voltage drop while simultaneously acting as a short-circuit with regards to the current (almost exactly like a diode). When the tube was powered at 170V with all the digits off, the voltages at the cathode pins was measured to all be around ~44V. When I bumped the supply voltage to 200V, the voltages changed to ~74V. This happens to be very close to (supply voltage)(turn off voltage). With a single digit turned on, the voltages on the other cathode pins varied depending on the distance of the measured digit to the one that is on. For the digit right next to the one on (physically, not numerically), the measured voltage came out to be exactly the same as the turn-off voltage. The further away the digit, the lower the voltage.

Driving Nixie Tubes

Most Nixie tube projects use vintage ICs 7441, 7441A, 74141, or the K155ID1 for driving the tubes. The 74141 and K155ID1 in particular are commonly used as a single IC is required per tube. They also have a built in BCD-to-decimal decoder, making it so that only 4 control lines were required to drive all digits of a given tube. These chips operate as current-sink drivers, sitting between the tube cathode pins and ground. The downside to using these chips is that they are relatively hard to find and only come in through hole DIP packages which takes up a significant amount of space on the PCB. Fine for most projects, but not ideal for the compact setup that I have in mind.

One alternative to these chips would be to use something similar to a shift register. For my test setup, the driver would have to handle the turn-off voltage of ~124V (unless the supply voltage was more than double the turn-off voltage). The problem is that drivers capable of handling such high voltages via open collector or open drain outputs are rare or difficult to drive. I highly recommend EEVblog’s videos on searching for a suitable Nixie driver.

Voltage Clamping with Zener Diodes

Fortunately, there is a way to work around the high voltage requirement. By connecting a zener diode to each output of a lower voltage rated driver, the output voltage can be safely clamped (pre-biased) to a level that won’t damage the driver. The 74141 in fact uses zener diodes to clamp the output voltages at 55V. This results in a voltage drop of (supply voltage)(zener clamp voltage) across the tube when the driver is off, which will be lower than the ignition/sustaining voltage if the clamp voltage is high enough. There are ICs that have this feature built in, such as the ULN2003 or the SN75468.

One possible issue to using zener diodes is that it changes the ‘current density’ inside the tube as a small amount of current is always drawn by each cathode. This results in an extra glowing ‘blue dots’ if the current density is too high as described quite well on this site. Details regarding this phenomenon is sparse, but it sounds like the issue can be avoided by clamping the output voltage to a higher level.

Switching with Transistors

There is a way to driving Nixie tubes without the use of zener diodes however, by opting for transistors that are capable of handling 200V or more. Modern bipolar transistors can easily meet this requirement while also being available in incredibly small packages: the MSD42WT1G at 10¢ each (quantities of 100+) is capable of switching up to 300V with a 2×2mm SOT-323 footprint. Combined with a logic level shift register, these BJT transistors can be configured as a simple switch to turn each digit of the tube on or off.

The downside to using this BJT and shift register combination is that it does not allow you to control the brightness of individual digits. This feature seems to be rare among other clock designs as most choose to dim the display by PWMing the anode supply. This works as a Nixie tube powered on behaves very much like an LED: to control the brightness of individual digits, the current or the PWM duty cycle for the each digit needs to be configurable.

Programmable Current Sink with BJTs

Similar to an LED, the brightness can be controlled by sinking a fixed current through each digit. One of the benefits of using a BJT transistor is that it can act as a fairly accurate current source in the right configuration. By varying the voltage at the base and placing a resistor between the emitter and ground, the collector current can be easily controlled. For example, a 1K ohm resistor and VB = 5V will result in VE = ~4.4V and IC = ~4.4mA regardless of VC. Given that VE and IC scales (mostly linearly) with VB, a few DACs with multiple outputs can be used to set the current for each individual digit between 0 and ~4.4mA on a 5V rail. This method introduces a few new issues however, the first being that each transistor now dissipates a significant amount of power (0.2W at 40V/5mA) when on. Another issue is that Nixie tubes have a lower limit for the operating current; the digits have uneven brightness at low currents, and cease to be fully lit when a threshold of ~0.5mA is reached:

Digit 0Digit 1Digit 2Digit 3Digit 4Digit 5Digit 6Digit 7Digit 8Digit 9

High Frequency Switching with BJTs

Again, similar to an LED, an alternative method for brightness control is to vary the on-period of each digit via a PWM signal. This can be done fairly easily by replacing the shift register with an IC that has configurable PWM outputs. One possible issue with this method of driving Nixie tubes in this fashion is that the display may exhibit visible flickering if the PWM frequency is low. Another possible issue with the PWM method is that the anode power supply may struggle to maintain a consistent voltage if the PWM period is synchronous (switching on/off at the same time) among all the drivers. If this happens, the supply current will rapidly alternate between zero and the desired current, which may result in extra instability from the boost supply switching between CCM and DCM modes.

One important difference between LEDs and Nixie tubes is that LEDs have near-instantaneous turn-on times while Nixie tubes have an inherent ignition delay. Because of this, there is a minimum on-time that must be met by the duty cycle. This is easily done by decreasing the modulation frequency, but comes at the cost of visible flickering. At higher frequencies, a longer duty cycle is needed to meet the minimum turn-on time, thus resulting in a higher minimum brightness. Ignition time is also reduced with higher anode voltages, potentially allowing for a lower minimum duty cycle.

Another important thing to take into consideration when choosing among the different method of driving Nixie tubes is the PCB footprint required for the driving circuitry. Discrete transistors will take up significantly more space compared to a single 74141, and even more so when additional ICs are required for driving the transistors. With a minimum of 40 transistors, 40 resistors, and a few ICs, the PCB becomes significantly more complex to route. There are ways to minimize the footprint however, by using resistor arrays and ICs like the DAC088S085.

Custom Driver PCB

After the initial testing was done, I went ahead and designed the first of two PCBs. The first PCB, the top half, contains only the components required for driving the Nixie tubes. Given the testing results, I opted to use discrete transistors and a few cascading PWM ICs to drive them. The PWM chips chosen was an expensive but feature-rich chip: the NXP PCA9685. With 12 bits per each of 16 outputs and an I2C interface, only three are needed to drive all 41 pins with some room to spare for future upgrades. There were three main features that I was looking for in the driver chip: 1) output driver capability, 2) large operating voltage range, and 3) ability to phase shift each output.

The PCA9685 fits the bill perfectly as the outputs can be configured as open drain or totem pole (most PWM chips for driving LEDs are open drain only). It also has an operating voltage range of 2.3V to 5.5V, so by changing the supply voltage, the baseline current is programmable from ~1.7mA to ~4.9mA. Finally, the delay between the rising edge of each output can be individually configured. With a bit of extra logic, the current consumption of the tubes can be load balanced to maintain a more consistent load on the power supply. By adjusting the phase of each output such that the ‘on’ period has minimal overlap, the peak current can be reduced by up to 75% compared to a driver with non-phase shifted outputs. This amount of savings only happens when the duty cycle is less than 25%, but otherwise it scales somewhat linearly to the point of maximum peak current at 100% duty cycle.

The other half of driving the outputs is the high voltage transistors. I opted to use discrete high voltage NPN transistors here: the ON MSD42WT1G. The main feature of this transistor is it’s voltage rating: with a maximum VCE and VCB rating of 300V, there is an additional 50% buffer for voltage transients. It also comes in one of the smallest SMD footprints available for transistors, the SOT-323. This is quite important due to the number of transistors needed. The other parameters are nice as well, with VBE of 6V being higher than the operating voltage of the PWM chips along and the capability to continuously handle up to 3mA at VCE of 200V.

Revision A

A significant amount of effort was put into routing the board to make the PCB as small as possible while only using two copper layers to keep the cost down. Routing a four layer board would have made things easier, but would have also doubled the cost. The resulting 20mm × 100mm PCB is just slightly larger than the bounding box of the tubes themselves when placed side by side. High and low voltage traces were also isolated as much as possible give the physical constraints of the board. High voltage traces (right input) are primarily routed on the top layer while logic level traces (left input) are routed on the bottom layer with both circuits having isolated grounds. Two holes for M2.5 screws were also placed in the middle for structural support using standoffs between the boards.

Revision B

There were a few issues that I wanted to address with the second revision of the driver board:

  • No ground shielding plane between the high voltage switching signals and the logic level signals.
  • Corners of the PCB needs to be rounded due to enclosure manufacturing requirements.
  • High voltage and logic grounds should be merged as the same ground reference is used by the driver (for current control) and by the Nixie tubes (cathode reference).
  • More structural rigidity from the signal/power headers.

To address these concerns while keeping the overall board dimensions the same, I had to re-design the board using four layers. The top layer is mainly used for components and high voltage switching traces. The following layer is mainly used as a ground and power plane. The bottom two layers are used for signal traces. The two additional layers in this revision both decreased trace routing complexity as well as added space for new features in the future.

Bill of Materials

Count Description Footprint Part Number
41 Transistor, NPN, 300V, 150mA SOT-323 MSD42WT1GOSCT-ND
41 Resistor, 1kΩ, 1/10W, 1% 0603 311-1.00KHRCT-ND
40 Sockets, 1mm   (Ebay)
3 PWM controller, PCA9685 QFN-28 568-5305-1-ND
1 Connector, two terminal spring cage   277-2086-1-ND
2 2×3 Pin header, male, 2mm pitch (revision B only)   H10183-ND
2 2×3 Pin header, female, 2mm pitch (revision B only)   H2851-ND

Nixie Power Supply

The other half to driving Nixie tubes involves the high voltage power supply. A high efficiency DC-DC boost converter circuit is commonly used in Nixie tube projects to generate such high voltages. There are two variations of this converter circuit: one uses a inductor (boost) while the other uses a transformer choke (flyback). Each has its own merits: inductors are usually available in larger selection of both value and footprints while transformer can utilize a winding ratio to reduce the peak current and voltage load on both the inductor and switch.

If you are looking for a good off-the-shelf power supply, YanZeyuan’s NCH6100HV is a pretty good choice. As a word of warning, don’t get the cheaper knockoffs as they use inferior components. The fake one I got was capable of sourcing ~200V but could only source 8mA compared to the 40mA from the NCH6100HV. The NCH8200HV is also a good choice if you are looking for a smaller high efficiency (86% @ 17mm × 18mm!) power supply. An example of a flyback design with minimal footprint for boosting 5V to 170V can be found here.

Power Supply Design

Rather than buy a commercial unit and integrating it into my project, I wanted to get some experience with the process of designing a power supply from scratch. The section below details the results of my investigations into power supply design as well as to provide a guide for those looking to build one of their own.

The two stages of operation for boost converters
The two stages of operation for boost converters

There are a number of excellent resources on the design and operation of boost converters, the foremost being the books Art of Electronics, Switching Power Supply Design, and Switchmode Power Supply Handbook. For web-based resources, this site covers the basics while these TI app notes go into more detail. This project page details the design of a custom high voltage boost converter, while this one does a good job at explaining how to optimize component placement on the PCB. There are similar articles and resources detailing the differences between boost and flyback circuits.

Disclaimer: while many of the equations below for boost components were directly copied from various app notes, most of the equations for flyback components were manually derived due to lack of references. See this page for more info.

SMPS Fundamentals: CCM vs DCM

CCM waveforms
CCM waveforms (Source: TI’s app note)

There are two fundamentally different operating modes for both boost and flyback converters. The first, continuous-conduction mode (CCM), occurs when the current in the inductor continues flowing during all stages of operation. The energy stored in the inductor increases while the switch is on and is equal to the energy discharged into the load through the diode when the switch is off, thereby ensuring steady-state operation. Because the current in the inductor never reaches zero, residual energy remains in the inductor at the end of each switching cycle.

DCM waveforms
DCM waveforms (Source: TI’s app note)

In discontinuous-conduction mode (DCM), the energy stored in the inductor when the switch is on is equal to the energy required by the load for one switching cycle. The energy stored in the inductor is depleted to zero when the switch is turned off, and results in a period where there is no current flowing through the inductor. This results in higher peak currents through the components as well as a larger ripple voltage at the output of the converter. The benefit of DCM operation is the lower inductance requirement which can lead to a smaller footprint for the inductor.

Selecting a mode of operation is essential in the design of a boost converter as it impacts both the component choice as well as the resulting performance and efficiency of the converter when operating under design constraints. Efficiency is lower in DCM mode as peak currents are higher (due to the faster/complete discharge of the inductor) through the components. Maintaining operation in CCM mode requires a minimum load current however, which may not be achievable if PWM is used to control the brightness of the Nixie tubes.

Inductance cutoff points as a function of input voltage and load current at a switching frequency of 50kHz
Inductance cutoff points for a boost circuit as a function of input voltage and load current at a switching frequency of 50kHz. (Credit: Jan Rychter)

Designing a converter that operates in CCM mode is also more difficult due to issues such as sub-harmonic instability and transitions between DCM (startup) and CCM (steady-state). Details of such issues are left out of this writeup, but can be found in resources listed at the start of this section.

Since I’m planning on utilizing PWM for brightness control with a focus on minimizing the overall footprint, designing my power supply to operate solely in DCM seems to be ideal. For both types of circuits, the duty cycle boundary between CCM and DCM should first be computed:

D_{ccm(boost)}=1-\dfrac{V_{in}}{V_{out} + V_D}

D_{ccm(flyback)}=\dfrac{V_{out} + V_D}{(N_{s:p}\times V_{in}) + V_{out}+V_D}

Where VD is the voltage drop across the rectifier diode and Ns:p is the secondary:primary turn ratio for the transformer. The small difference in duty cycle when N=1 is due to the minimum voltage across the inductor: in a boost circuit, the voltage drops to Vin while the voltage across the secondary winding of a flyback’s transformer drops to zero. To compute the boundary value for the inductance required for CCM or DCM:

L_{(boost)}=\dfrac{V_{in}}{2 \times I_{out}\times F_S}\times D_{ccm}\times (1-D_{ccm}) and L_{(flyback)}=\dfrac{V_{in}^2\times D_{ccm}^2}{2\times I_{out}\times (V_{out}+V_D)\times F_S}

Where FS is the switching frequency. As depicted in the graphs above, the converter will operate in DCM if the inductance is kept below the the boundary value.

While the operating duty cycle (inductor charging/discharging) in CCM is dependent solely on the input and output voltages, the same does not apply in DCM as there is a period where the inductor is completely discharged. Furthermore, in DCM, the duty cycle varies more with load changes than it does when running in CCM. Many boost controllers have both a maximum and minimum switching frequency so care must be taken to keep the operating duty cycle within the capabilities of the boost controller. The DCM duty cycle can be computed by taking the inductance and switching frequency into account:

D_{dcm(boost)}=\sqrt{\dfrac{2\times L\times (V_{out}+V_D-V_{in})\times I_{out}\times F_S}{V_{in}^2}} and D_{dcm(flyback)}=\sqrt{\dfrac{2\times L\times (V_{out}+V_D)\times I_{out}\times F_S}{V_{in}^2}}

Ddcm here is the duty cycle at the given switching frequency when the switch is on (inductor is charging). The duty cycle for the discharge period can be found with:

D_{dis(boost)}=\dfrac{V_{in}}{V_{out}+V_D-V_{in}}\times D_{dcm} and D_{dis(flyback)}=\dfrac{V_{in}\times N_{s:p}}{V_{out}+V_D}\times D_{dcm}

The time in which the inductor is active is D_{dcm}+D_{dis}, and should be taken into consideration when choosing the switching frequency. Leaving an additional 20% buffer is recommended to ensure that the circuit operates in DCM.


Peak inductor current as a function of inductance and output current at a switching frequency of 50kHz from a 12V source
Peak inductor current as a function of inductance and output current at a switching frequency of 50kHz from a 12V source (Credit: Jan Rychter)

There are two main considerations that should be taken when choosing an inductor for a boost converter: the inductance and maximum current rating. Additional considerations include the winding resistance, shielding, core material, and size. For transformers, the turn ratio and the leakage inductance should also be taken into account. The (primary) inductance value needs to be less than the CCM threshold in order to maintain DCM at the desired load.

The maximum current rating is also important as the peak current through the inductor will be significant higher than the average load current. The peak current in the inductor or primary winding of a transformer can be computed as follows:

I_{pk(pri)}=\dfrac{V_{in}\times D_{dcm}}{L\times F_S} (in DCM only)

Where L is the inductance of the primary coil of the transformer for a flyback circuit and I_{pk(sec)}=I_{pk(pri)}/N_{s:p}. Due to this high ripple current, when selecting an inductor, a core material with low permeability and low core losses at the switching frequency should be chosen to minimize power dissipation. A shielded inductor will result in less RFI but will also slightly lower the overall efficiency. Care should also be taken to ensure that the peak current does not exceed the saturation current rating of the inductor or transformer. Conduction losses in the inductor for a boost circuit can be approximated with:

Pd_{L(cond)}=I^2_{L(rms)}\times R_L where I_{L(rms)}=\sqrt{I^2_{S(rms)}+I^2_{D(rms)}} (boost topology, in DCM only)

For flyback circuits, conduction losses are calculated separately for the primary and secondary windings:

Pd_{L(cond)}=(I^2_{S(rms)}\times R_{L(pri)})+(I^2_{D(rms)}\times R_{L(sec)}) (flyback topology, in DCM only)


CCM vs DCM waveforms
CCM vs DCM waveforms (Source: TI’s app note)

There are four important parameters when selecting a MOSFET switch: VDS (drain-source breakdown voltage), RDS(on) (drain-source on resistance), Qg (gate charge), and Coss (output capacitance). The breakdown voltage VDS should be greater than or equal to 1.5x the expected voltage across the switch. For a boost circuit, VDS(max) = Vout. For a flyback circuit, VDS(max) = Vin + Vclamp where Vclamp is the clamp circuit voltage. Design guidelines for snubber and clamp circuits is provided later in this section.

Power dissipation of the switch is comprised of gate, conduction, and switching losses. The gate losses can be approximated using the gate capacitance, gate voltage and switching frequency:

Pd_{S(gate)}=Q_g\times V_G\times F_S

The on-resistance of the switch, RDS, is used to approximate the conduction losses:

Pd_{S(cond)}=I^2_{S(rms)}\times R_{DS} where I_{S(rms)}=I_{pk}\times \sqrt{\dfrac{D_{dcm}}{3}} and I_{S(avg)}=I_{pk}\times \dfrac{D_{dcm}}{2} (in DCM only)

Switching losses originate from having to charge and discharge the node capacitances (Coss and Crss) along with losses during the transition periods from the Miller effect at low gate voltages:

Pd_{S(switch)}=\dfrac{F_S}{2}\times\bigg[C_{oss}\times\bigg(\dfrac{V_{out}+V_D}{N_{s:p}}\bigg)^2+\bigg(\dfrac{V_{out}+V_D}{N_{s:p}}\bigg)\times \dfrac{I_{out}\times N_{s:p}}{D_{dis}}\times\dfrac{Q_{gd}\times R_G}{V_G-V_{G(th)}}\bigg] (in DCM only)

Where N is 1 for an inductor (boost circuit). For a more through explanation regarding power losses in switching MOSFETs, reference these app notes. Note that the switching losses needs to be doubled as there are two charge/discharge cycles per switching period.

Rectifier Diode

The output rectifier diode must be capable of handling the peak reverse voltage, peak input current, and the diode’s average power. In a boost circuit, V_{D(rev)} > V_{out}. In a flyback circuit, V_{D(rev)} > N_{s:p}\times V_{in} + V_{out}. A simple approximation of conduction losses in the diode is:

Pd_{D(cond)}=I_{out}\times V_D where I_{D(rms)}=I_{pk}\times \sqrt{\dfrac{D_{dis}}{3}} and I_{D(avg)}=I_{pk}\times \dfrac{D_{dis}}{2} (in DCM only)

For lower output voltages, a low capacitance Schottky diode is recommended. For higher output voltages, a fast recovery diode should be chosen. Soft recovery characteristics, normally desirable when operating in CCM, are not required when operating in DCM as there is no current flowing through the diode when the switch is on. Switching losses associated with a Schottky diode is based off the diode’s capacitance while the losses when using a fast-recovery diode is determined by the recovery time:

Pd_{S(schottky)}=\dfrac{1}{2}\times C_{D}\times F_S\times(V_{out}+V_D)^2 and Pd_{S(rect)}=\dfrac{1}{2}\times V_{out}\times I_{pk}\times t_{rr}\times F_S (in DCM only)

For higher efficiency in low-current applications with high output currents or low output voltages, it is often advantageous to implement synchronous rectification with a second MOSFET in place of the diode. This adds additional implementation complexity, but greatly increases efficiency as the voltage drop across the rectifier diode is effectively negated. Some controllers even have built-in support for a synchronous MOSFET. Reference this TI app note for more information.

Synchronous rectification with a MOSFET does not really work with high output voltages however. The reason for this is because the gate-source voltage of the MOSFET needs to be higher than peak output voltage in order to turn the MOSFET on. Needless to say, generating a gate voltage higher than the output voltage is quite tricky to do.

Input/Output Capacitors

Capacitors are required in order to limit the voltage ripple at both the input and output. In a flyback topology, the input current is discontinuous so more capacitance is typically needed. Ceramic capacitors (with negligible ESR) are typically chosen to limit the amount of ripple voltage while tantalum capacitors are typically used to add bulk input capacitance to prevent large voltage swings when switching high loads. In both cases, the ripple current rating of the capacitor must be greater than IS(rms) (input) or ID(rms) (output).

The voltage drop across the capacitor can be computed as a function of the current, capacitance, and time: \Delta V=(I \times∇\Delta t)/C. Assuming a steady-state load current, the minimum capacitance and maximum ESR of the input capacitor to maintain a given voltage ripple at the input can be estimated as follows:

C_{in(cap)}>\dfrac{I_{S(rms)}\times (1-D_{dcm})}{V_{ripple(in)}\times F_S} and C_{in(esr)}<\dfrac{V_{ripple(in)}}{I_{S(pk)}} (in DCM only)

Capacitor current flow
Capacitor current flow

The output capacitor is used to reduce the ripple voltage at the output. The minimum capacitance and maximum ESR of the capacitor to maintain a given voltage ripple at the output can be estimated as follows:

C_{out(cap)}>\dfrac{I_{D(rms)}\times (1-D_{dis})}{V_{ripple(out)}\times F_S} and C_{in(esr)}<\dfrac{V_{ripple(out)}}{I_{D(pk)}} (in DCM only)

Power dissipation of the output capacitor can be computed using the current and capacitor’s ESR:

Pd_C=I^2_{C(rms)}\times R_{esr} where I_{C(rms)}=I_{D(rms)} (in DCM only)

Snubber and Clamp Circuits

Boost power stage with parasitic elements
RC snubber for a boost power stage with parasitic elements (Source: TI’s app note)

In addition to the main components listed above, some additional parts are needed to ensure stability in various parts of the circuit. Parasitic components such as inductor shunt capacitance and lead-wire inductance in the MOSFET and diode can cause ringing at the switch node in excess of 100Mhz. This ringing generate EMI and will damage the MOSFET if the ringing voltage peak is too high.

To address this issue, a RC snubber can be added in parallel to the MOSFET to dampen the parasitic energy. Methods to optimize the RC values is described in these app notes. The initial RC values can be estimated with:

C_{sn}=\dfrac{I_{C(pk)}\times t_r}{V_{C(max)}} and R_{sn}=\dfrac{t_{on(min)}}{3\times C}

Where tr is the desired rise time at the switch node and VC(max) is the maximum voltage across the MOSFET. The power dissipation of the snubber resistor can be approximated with:

Pd_{R(sn)} = \dfrac{1}{2}\times C_{sn} \times V_{sn(max)}^2\times F_S

This ringing is especially problematic in flyback transformers due to the leakage inductance between the transformer windings. The leakage energy resulting from the leakage inductance is kept in the primary winding when the switch is turned off, resulting in a large voltage spike at the MOSFET’s drain terminal.

Snubber circuit for leakage inductance
RCD clamp circuit for leakage inductance (Source: TI’s app note)

To protect the MOSFET from this voltage spike, a RCD clamp circuit should be added in parallel across the primary winding of the transformer to create a low impedance voltage source. This circuit clamps to a voltage that is usually 2x higher than the reflected output voltage (Vout/Ns:p) to keep the voltage across the MOSFET within an acceptable range. A higher clamp voltage decreases the switching time (and thus losses), but increases EMI and MOSFET drain voltage.

This voltage is relatively constant during a switching cycle and reaches its maximum value when powering a full load with minimum input voltage:

R_{clamp}=\dfrac{2\times\bigg(V_{clamp}-\dfrac{V_{out}}{N_{s:p}}\bigg)\times V_{clamp}}{F_S\times L_{leak}\times I_{pk}^2} and C_{clamp}=\dfrac{V_{clamp}}{V_{clamp(ripple)}\times R_{clamp}\times F_S}

Where Lleak is the leakage inductance on the primary winding, Ipk is the peak current on the primary winding, Vclamp is the desired voltage to clamp the MOSFET’s drain to, and Vclamp(ripple) is the voltage ripple across Cclamp (5-10% of Vclamp). The diode should have a fast forward recovery time to reduce the voltage spike as well as being able to handle Ipk with a reverse voltage rating greater than Vclamp + Vin. The power dissipation of the clamp circuit can be approximated with:

P_{clamp}=\dfrac{1}{2}\times L_{leak}\times I_{pk}^2\times F_S\times \dfrac{V_{clamp}}{V_{clamp}-N_{s:p}\times (V_{out}+V_F)}

DZ clamp circuit for leakage inductance
DZ clamp circuit for leakage inductance (Source: Linear’s app note)

An alternative to the RCD clamp is the DZ (diode-Zener) clamp. Schottky diodes are typically the better choice here due to their faster turn-on time. Similar to the RCD clamp, the diode needs to be able to handle Ipk and have a reverse voltage greater than Vclamp + Vin where Vclamp is set by the Zener reverse breakdown voltage. The power loss in the clamp circuit determines the required power rating of the Zener diode. Power loss is highest maximum load and minimum input voltage. A higher Zener breakdown voltage will reduce the power loss but the breakdown voltage rating must be below the MOSFET’s voltage rating (VDS(max)Vin) with some margin.

P_{clamp}=\dfrac{1}{2}\times L_{leak}\times I_{pk}^2\times F_S\times \bigg[1+\dfrac{V_{out}+V_F}{N_{s:p}\times V_{zener}-(V_{out}+V_F)}\bigg]

Where Vzener is the Zener diode breakdown voltage, VF is the voltage drop across the clamp diode and VD is the voltage drop across the output rectifier diode. The DZ clamp should have slightly higher efficiency compard to the RCD clamp circuit at the cost of higher EMI.

While a clamp circuit is typically required, omitting the clamp or snubber is possible if the peak voltage resulting from the transformer’s leakage inductance is lower than the switching MOSFET’s voltage rating. This is best determined by empirical means, but can also be estimated with:

V_{pk(pri)}=I_{pk}\times \sqrt{\dfrac{L_{leak(pri)}}{C_L+C_{oss}}}+V_{in}+\dfrac{V_{out}}{N_{s:p}}

Where Vpk is the peak voltage seen at the drain of the MOSFET, Lleak(pri) is the primary leakage inductance, CL is the transformer capacitance on the primary winding, and Coss is the output capacitance of the MOSFET. The leakage inductance on the secondary winding also results in some ringing when coupled with the reverse recovery current of the output rectifier diode. This overshoot may result in significant EMI as well as possibly exceeding the rectifier diode’s voltage rating. The voltage peak on the secondary can be estimated with:

V_{pk(sec)} = I_{rev}\times \sqrt{\dfrac{L_{leak(sec)}}{C_D}}}+V_{in}\times N_{s:p}

Where Irev is the reverse recovery current, Lleak(sec) is the leakage inductance on the secondary winding, and CD is the rectifier diode’s capacitance. Additional margin should also be added to the peak values to account for parasitics.

SMPS Controller IC

Last but not least, there is the controller which contains the logic and circuitry for driving the MOSFET and regulating the duty cycle and/or switching frequency to maintain the desired output voltage at any given load. These controller ICs can range from simple 555 timers to feature-rich integrated controllers such as the MAX1771, TPS40210, or LT8304. There are many more controller available from a variety of vendors for DC-DC converters with various features that separate them from others:

  • Pulse-frequency modulation (PFM) vs pulse-width modulation (PWM)
  • Integrated MOSFET or MOSFET drivers
  • Slope compensation (for CCM)
  • Soft-start capability
  • Switching frequency ranges from 20kHz to 1MHz
  • Overcurrent detection/shutdown
  • Maximum operating power, voltage, and current ratings
  • Available footprints and sizes
  • Required number of external components

Custom Power Supply

The choice of components used is critical to both performance and lifespan of the boost converter. For this particular project, the focus was placed on creating a minimally sized flyback converter capable of driving a maximum load of 20mA at high (>90%) efficiency in DCM for fast transient response. The input to the converter will be sourced from a 12V wall wart along with a 3.7V lithium polymer battery for standalone operation. A 3.3V or 5V voltage rail for the microcontroller will generated using a buck/boost converter. The output will be fixed at 200V.

To get a better understanding of the various parameters involved in component selection, I created a few spreadsheets to solve for various parameters (Ddcm, Ddis, Ipk, Irms, PdS, PdL, and PdD) of the circuit across a range of inductance and switching frequencies for both boost and flyback topologies. From these sheets, it’s easy to see that a flyback circuit has lower overall power loss in the inductor(s), switch, and rectifier.

Transformer Selection

For the transformer, I filtered the options available on Digikey for surface mounted transformers between 8mm and 11mm in height. Anything smaller had either too high of an inductance value, low winding ratio, or a low saturation current. I took the options that were available and plugged them into another page in the spreadsheet to compute the duty cycles and estimated power loss. Note that the choice of MOSFET and rectifier were kept the same across all options, even though the MOSFET will likely change depending on the winding ratio of the transformer.

From the results, I looked at a few different combinations of input voltage and output current to get an idea of the power consumption across the range of available transformers. With the minimum input voltage of 3.7V and an output current ranging from 1mA to 25mA, the Wurth 750315825 or 750312367 seems to provide optimal performance when operating at 100kHz. Note that these transformers were originally designed for buck converters, so the windings and their corresponding ratings need to be swapped. Of the two transformers, the 750315825 would have been a better choice due for its lower inductance (which allows for higher output currents), but that part doesn’t seem to exist outside of Wurth’s catalog.

The saturation current rating of the transformer was anothe point of concern given the high peak current through the primary winding of the transformer. To verify that the performance of the transformers, LTSpice simulation was done using the models provided by the manufacturer to verify that the true saturation point (where the current starts to sharply rise) is much higher than the rated saturation current.

TIND = Wurth 750312367
Rated inductance (primary): 4.25µH
Winding ratio: 1:8.4
Resistance (primary): 30mΩ
Resistance (secondary): 800mΩ
Saturation current (primary): 3.8A
Leakage inductance: 1.7µH
Footprint (L×W×H): 13.4mm × 10.2mm × 8.7mm

MOSFET Selection

Once the transformer has been picked out, the next step is to choose the switching MOSFET. The expected voltage across the MOSFET is Vin + Vclamp where Vclamp is double the reflected voltage of 24V. This sets a minimum requirement of VDS = 60V. The other important parameters are the on-resistance RDS, output capacitance Coss, and the overall footprint size. Note that the datasheet must be thoroughly examined if the gate drive voltage is close to the threshold voltage as the drain-source resistance will increase exponentially when the gate voltage is low.

The 3.7V input requirement makes it difficult to reduce switching losses at the minimum input voltage. Most MOSFETs, even those with logic level gate thresholds, will exhibit extremely high drain-source resistance at 3.7V. For example, the On-Semi FDMC86160 has a drain resistance of 16mΩ and a gate threshold of only 2.9V, but if you look at the RDS(on) to VGS chart in the datasheet, the drain resistance spikes to an unknown value that could potentially be in the tens or hundreds of ohms at low gate voltages! With an input voltage of 3.7V and an output of 25mA, the MOSFET will see a RMS current of 2.1A which will result in extremely high losses!

Another interesting observation when choosing MOSFETs is that you have to either prioritize the on-resistance or the output capacitance, but not both at the same time. MOSFETs with an extremely low on-resistance of <50mΩ typically have a much larger output capacitance of >300pF. These are a better choice for high output current applications where the conduction losses dominate. MOSFETs with a low capacitance of <300pF on the other hand typically have a larger on-resistance of >200mΩ. These are better for low output current applications where switching losses tends to dominate. This makes sense as a wider switching channel results in lower resistance, but adds more material and therefore more capacitance to the device.

Given that this is a one-off project with efficiency being heavily prioritized, I found a novel MOSFET available on Digikey: the EPC2016C. What makes this part unique is that it is not a traditional Silicon MOSFET, but a Gallium Nitride one that is a only available in passivated die form to keep the on-resistance and gate capacitance to an absolute minimum. The manufacture also provides LTspice models, detailed guides to using eGaN FETs, as well as assembly and layout guidelines for their parts.

QSW = EPC2016C
Max drain to source voltage: 100V
Drain to source resistance: 16mΩ @ 3.7V
Minimum gate voltage: 2.85V
Output capacitance: 210pF
Continuous current: 18A
Footprint: 3mm2

Rectifier Diode Selection

With the transformer known, the rectifier diode can be chosen. The peak reverse voltage across the diode includes the reflected voltage from the secondary: N{s:p} × Vin + Vout, resulting in a minimum reverse voltage of 300V.

Max reverse voltage: 400V
Forward voltage: 0.85V @ 200mA
Reverse recovery time: 20ns
Footprint: 8mm2

Input/Output Capacitor Selection

Choosing the input and output capacitors were mainly driven by what was available than a desire for a specific ripple voltage. Ceramic capacitors were chosen for each end: one for bulk capacitance and one for high frequency transients. Capacitors with a voltage rating higher than the required voltage was selected to account for the capacitance drop due to DC bias. This mostly impacted the output capacitors as the 2.2μF capacitor is effectively a 1.43μF capacitor at 200V while the 100nF capacitor turns into a 60nF capacitor at 200V.

Input Capacitors:
CIN (bulk input) =445-14933-1-ND (10µF ± 20%, 50V, X7R, 1210)
CIN (HF bypass) = 445-7856-1-ND (100nF ± 10%, 50V, X7R, 0805)

Output Capacitors:
COUT (bulk input) = 445-7781-1-ND (2.2µF ± 10%, 450V, X6S, 2220)
COUT (HF bypass) = 445-7922-1-ND (100nF ± 10%, 630V, X7R, 1812)

Clamp Component Selection

A DZ clamp is used to protect the MOSFET from the leakage transformer’s leakage inductance. The diode needs to have a reverse voltage rating greater than Vin + Vclamp = 60V as well as being able to handle the peak current of 2.1A. The Zener diode’s reverse breakdown voltage must be below 88V with some extra margin to ensure that the voltage rating of the MOSFET is not exceeded. The Zener diode must also be able to handle a peak load of 625mW. Diode selection was limited by the selection of available footprints, so I just went with the smallest footprint.

DSN = V2PM10HM3/H (100V, 2A, Schottky)
DZSN = BZD27C75P-E3-08 (75V, 800mW)

Components for the clamp circuit was selected to serve as a backup in the event that the the peak voltage did exceed the expected maximum of 90V with the 750312367 as the transformer and the EPC2016C as the switch. If the measured voltage spike remains under the 100V rated limit at maximum output current, the clamp circuit will be omitted from the final design. A snubber circuit across the MOSFET and output rectifier diode was omitted to minimize component count and board size as reducing EMI is not a high priority for this project.

Boost Controller Selection

For this project, I decided to go with the LT3759 for the boost controller. This chip was chosen in particular for its wide operating voltage range from 1.6V to 42V with a regulated gate voltage of either 3.75V (from VIN) or 4.75 (from VDRIVE). A secondary input for the internal voltage regulator from the boost converter’s output allows for a higher gate drive voltage to reduce the switch MOSFET’s on-resistance.

With the EPC2016C, a gate voltage under 2.85V causes a dramatic rise in the drain to source resistance. This is acceptable as lithium-ion batteries operate from 3.0V to ~4.0V and the internal regulator of the LT3786 has a 0.15V drop between Vin and INTVCC when Vin is under 3.9V.

The auxiliary components for the LT3759 can be selected based off of formulas given in the datasheet:
RSENSE = 10mΩ (peak 5A on primary)
RFREQ = 86.6kΩ (for 100kHz)
CSS = 150nF (18ms soft start)
RVC = 300kΩ (from simulation)
CVC = 560pF (from simulation)
RFB = 1.6MΩ (120nA FBX pin input current)
RBIAS = 13kΩ (1.6V at FBX pin)
CBP = 4.7µF (from datasheet)

For lower output currents, the TPS40210 is an interesting alternative with its lower rated operating frequency of 35kHz. For a design optimized for overall footprint, the LT8304 has a built-in 2A 150V switch for a larger reduction in component count, with the trade-off being somewhat lower efficiency.

LTspice Simulation

Prior to finalizing the components and laying out the PCB, some simulation was done in LTspice to verify that the circuit operated as expected in a somewhat ideal scenario. Models for the transformer and the boost controller were provided by the manufacturer, and similar components were chosen from the LTspice component library for the MOSFET switch and the rectifier diode. Some important points that were observed:

  • The transformer (750312367) fully saturates at a peak of 5A. This limits the output current to a maximum of 18mA regardless of the input voltage.
  • Make sure the transformer is using the saturation model rather than the default linear model.
  • A fast recovery diode for the output rectifier is absolutely necessary. The reverse recovery time is a major factor in how much energy is transferred to the output capacitors.
  • No significant voltage spike or ringing was observed on the switching node.

Design Specifications

Parameter Symbol Test Conditions Min Typ Max Unit
Input Voltage Vin  Ambient = 25°C 3.0 3.7 12 V
Input Current Iin Vin = 3.7V, Iout < 25mA, η = 90%   1.17 2.33 A
Vin = 12V, Iout < 25mA, η = 90%   0.65 1.29
Output Voltage Vout RFB = 1.6MΩ, RBIAS = 13kΩ   200 205 V
Output Current Iout     10  18 mA
Output Overcurrent Iout(max) RSENSE = 10mΩ     20 mA
Switching Frequency FS RFREQ = 86.6kΩ 100 kHz
Typical Efficiency ηpk Vin = 3.7V, Iout = 10mA > 90 %
Vin = 12V, Iout = 10mA > 90
Full Load Efficiency η Vin = 3.7V, Iout = 25mA > 85 %
Vin = 12V, Iout = 15mA > 85
Operating Temperature Range Top   -20 25 125 °C


Bill of Material

Designator Description Footprint Part Number
C1 CIN – Ceramic, 10µF ± 20%, 50V, X7R 1210 445-14933-1-ND
C2 CIN – Ceramic, 100nF ± 10%, 50V, X7R 0805 445-7856-1-ND
C3 CVC – Ceramic, 560pF ± 10%, 50V, X7R 0603 445-5649-1-ND
C4 CSS – Ceramic, 150nF ± 10%, 50V, X7R 0603 445-6925-1-ND
C5 CBP – Ceramic, 4.7µF ± 10%, 35V, X5R 0603 445-9064-1-ND
C6 COUT – Ceramic, 2.2µF ± 10%, 450V, X6S 2220 445-7781-1-ND
C7 COUT – Ceramic, 100nF ± 10%, 630V, X7R 1812 445-7922-1-ND
D1 DZSN – Zener, 75V, 0.8W DO-219AB BZD27C75P-E3-08GICT-ND
D2 DSN – Schottky, 100V, 2A MicroSMP V1PM10-M3/HGICT-ND
Q1 QSW – MOSFET, N-channel, 100V, 16mΩ 2.1mm × 1.4mm 917-1080-1-ND
R1 RFREQ – Resistor, 86.6kΩ, 1/10W, 1% 0603 P86.6KDBCT-ND
R2 RVC – Resistor, 300kΩ, 1/10W, 1% 0603 P300KHCT-ND
R3 RBIAS – Resistor, 13kΩ, 1/10W, 0.1% 0603 P13.0KHCT-ND
R4 RSENSE – Resistor, 10mΩ, 1/2W, 1% 0805 P.01AQCT-ND
R5 RFB – Resistor, 1.6MΩ, 1/10W, 1% 0603 P1.60MHCT-ND
T1 TIND – Transformer, 4.25µH, 1:8.4 13.4mm × 10.2mm 1297-1017-1-ND
U1 Boost Controller, LT3759 MSOP-12 LT3759EMSE#TRPBFCT-ND

PCB Layout (Rev. A)

The high speed operation of a boost converter PCB requires careful attention to board layout and component placement. To prevent EM radiation and high frequency coupling into signal lines, proper layout of all components on the PCB is necessary. This is especially true for components around paths that contain fast changes in current (high di/dt) to reduce inductive ringing. In a boost topology, this loop contains the output capacitor, the sense resistor, the MOSFET switch, and the rectifier diode. In a flyback topology, the primary loop contains the input capacitor, the primary winding of the transformer, the MOSFET switch, and the sense resistor. The secondary loop contains the output capacitor, the secondary winding, and the rectifier diode.

As usual, there are a number of excellent resources on this topic. The set of guidelines that I followed:

  • Use a Kelvin connection between the sense RSENSE and the SENSE pin
  • Keep high current paths away from the FBX pin
  • Place the divider resistors near the FBX pin to keep the high impedance FBX node small
  • Keep RVC and CVC near the VC pin for similar reasons
  • Bypass capacitors should be kept close to the IC
  • No ground plane under the inductor/transformer
  • Ground RSENSE with multiple vias to reduce parasitic resistance and inductance
  • Keep the trace from the GATE pin to the MOSFET short

Following these guidelines, I manged to lay out the components on a 29mm × 16mm PCB.

Design Analysis (Rev. A)

The PCB was assembled using an IR reflow oven and a stencil mask for OSHstencils. As with all boards that I make, there are a few notes on improvements that can be made for the next revision:

  • Having a correct solder mask for the EPC2016C is absolutely critical due to the pad sizes.
  • Under-sizing the solder mask is recommended as some solder paste will get underneath the stencil.
  • The transformer should be manually soldered to avoid melting the tape.

Once the board was assembled, I scoped a number of nodes on the PCB to find any potential issues with the design and layout. For the most part, my power supply look nominal. The output voltage was steady at 198V across the 3V to 12V input range. Most of the waveforms from the various nodes in the circuit looked correct as well, with the only exception being some unexpected ringing at the source node (voltage across the sense resistor). Originally I thought that I may have damaged the MOSFET from excessive heat when I had to fix a solder joint, but the issue persisted even after replacing the MOSFET. Some observations:

  • Ringing at the drain node peaked at ~50V with a peak of 5A in the primary winding.
  • The controller limited the current in the primary loop to 5A (50mA) as designed.
  • Gate drive was operating at the expected voltage across the input range.
  • MASSIVE ringing at the source node when the MOSFET switched on and off.
    • Ringing increased with higher input voltages.
    • Same measurement for the NCH8200HV was done as well for comparison.
  • Building a variable resistive load capable of sinking 5W out of discrete resistors is rather difficult to do.

After some discussions with folks who are more knowledgeable than me, it sounded like there were a few potential factors that could be contributing to the ringing. Of these, the largest factor was the lack of a resistor between the gate driver and the gate of the MOSFET. When the low gate-source capacitance is driven with a high current (~6A) from the controller IC, parasitic inductance in the traces and components will cause ringing to occur. Furthermore, the current loop between the IC ground and the ground of the sensor was rather large, leading to a bit of ground bounce. Finally, the controller IC should be bypassed with an additional (smaller) capacitor to provide some extra current boost for driving the gate of the MOSFET.

PCB Layout (Rev. B)

For the second revision of the board, I wanted to address the following concerns:

  • Add a resistor between the IC gate driver pin and the gate of the MOSFET.
  • Increase the number of vias connecting components to the ground plane.
  • Decrease the loop size between the gate driver, MOSFET source, and sense resistor.
  • Remove the clamping diodes as they are unnecessary for the peak current of this transformer.
  • Use OSHpark’s option for 0.8mm thickness with 2oz copper for the PCB.

Controller Board

Now that all concerns regarding the Nixie tubes have been answered, the last stage of the design (regarding the electronics components at least) is the controller board that ties everything together. The main focus of this board is the microcontroller, with additional (optional) auxiliary components being the real-time clock, audio driver and amplifier, lithium-ion battery charger, backlight LED driver, GPS, and wireless communication SoCs.


There were two main drivers behind the choice of microcontroller for this project. The first is that I wanted to use a chip/architecture that I have not used before in any of my previous projects. The second is to follow the open-source nature of this project and build upon a platform that has an open and easily available toolchain. For obvious reasons, I also wanted to use something that is at least somewhat popular/common in the industry to get some experience with working with such chips and architectures.